Wireless World, August 1973
Second of three articles describing the operation and construction of a modular system with
manual or electronic voltage control of synthesized waveforms.
By T.Orr B.Sc. and D. W. Thomas Ph.D., M.I.E.R.B
Please note: all copyrights by the authors and Wireless world.
Pictures and schematics can be found halfway this article.
Continuing the construction with descriptions of voltage control circuitry, reverberation and the exponental converter
The first part of this series of constructional articles (August issue) described the philosophy behind the design of the synthesizer and its capabilities as a musical or educational instrument. The series continues with constructional detail of the circuitry. Each basic modular unit is described in full, but the number of units employed can be varied to suit the constructor¥s needs
Sweep frequency oscillator
By driving VCO1 with a ramp, generated by VCO2 (both described last month), it is possible to produce a sweep frequency oscillator capable of covering the entire audio spectrum in one sweep (Fig. 14). If the swept sinewave output of VC01 is then fed into a network, the amplitude-frequency response of that network can be rapidly determined. A three decade sweep is available and the peak to peak amplitude is virtually constant. However, the sinewave generated by VC01 is by no means pure, having a harmonic content of between 3 and 4%. This limits the resolution to a rather modest value, but even so, a reasonable representation of the network’s frequency response can be obtained. (It is particularly useful for directly observing the effect of tone controls in audio amplifiers.) To display the amplitu˛e-frequency response, the ramp drives both the oscillator VCO1, and an oscilloscope (in the x-direction),whilst the network response is displayed in the y-direction. The drive need not be a ramp; in fact any continuous function could be used.
Voltage controlled oscillator, VCO3
This oscillator produces a sequence of steps, the amplitude of the steps being individually controllable. The number of steps in the sequence can be varied up to a maximum of six (Fig. 15) and a series of pulses is also available (1 to 1 mark/space ratio) each being coincident with the leading edge of each step. The oscillator is voltage controlled and has a pair of summing inputs.
The frequency-voltage relationship is exponential and extends from subsonic frequencies to above 20kHz, all in one range. The oscillator, which consists of a voltage controlled astable driving a binary counter, is shown in block diagram form in Fig. 16.
The b.c.d. output is decoded into decimal form, attenuated by pots 1 to 6 and then summed. The length of the resulting sequential output can be modified by selectively resetting the binary counter. The circuit diagram of VCO3 is given in Fig.17. Transistors Tr2 and Tr3 are voltage driven and provide a current drive to the astable. The result is an exponential current-voltage relationship and an exponential frequency-voltage response.
The useful range of the control voltage applied to Tr2,3 is a few hundred millivolts, and must be generated relative to +Vcc. Preset R7 is adjusted so that the clock frequency produced is approximately 2OkHz with the bias set at maximum. Also, when the bias is set to a minimum, preset R10 is adjusted so that the clock frequency is approximately 0.2Hz. Unfortunately the effects of R7 and R10 are interdependent and hence they must be set up iteratively until convergence is achieved on the desired settings.
The logic section is self explanatory ; however, the logic power supply must be decoupled with a 0.1uF capacitor. Note that this oscillator does not lend itself to construction on plug-in boards, as the edge connections required are exceptionally large.
Voltage controlled amplifier, VCA 1, 2
The heart of this unit which performs the function of amplitude modulation is a linear four quadrant multiplier, the device being an integrated circuit SG1495D. This device operates from a + 15V supply, and when used in the circuit shown in Fig.18 can accept inputs of +/-5V. The frequency response is greater than that required. The output is taken between two load resistors and a differential amplifier (IC2) is required to remove the common mode signal. Design of the multiplier and differential amplifier is very nearly the same as that given in the applications sheet for the SG1495D but some component values have been modified and lower tolerances are used. A scale factor of 0. 1 is employed.
The multiplier accepts two inputs, X and Y, and generates an output that is linearly proportional to the product XY. The X input is the audio signal VA1, and the Y input is the output of the control cinuit. This circuit is a voltage summer with inputs of VC1, VC2 and a bias voltage. This arrangement is that of a “perfect” halfwave rectifier, thus, when the sum of the control and bias signals goes negative, the output of IC3 remains at OV. Only when the sum is positive will an output (a control voltage9 be produced. In this way, the control section has a threshold characteristic, this threshold occurring at OV and being used as the reference level for zero
output from the v.c.a. When long leads an used, parasitic oscillation may occur, but this can be suppressed by an RC network (C1R2).
Aligning the VCA’s
Four presets (Fig. 18) have to be aligned ; these are R20, R21, R14, and R8. The first two are Y and X “offset adjust”, the third is “gain” and the last is “output offset”.
Set X and Y to 0V making sure that the bias (R25) is set at its most negative setting. Monitor the output of IC2 and adjust the output offset (R8) until it is at zero potential.
Set X to + 5V, but keep Y at 0V, and adjust R20, the Y “offset adjust” until the output is again at zero potential.
Set Y to +5V, set X to 0V, and adjust R21, the X “offset adjust”, until the output is once more at zero potential.
Now repeat the first step. The last preset to be adjusted is R14, the gain control, which alters the scale factor of the generated output. The unit is now a functional v.c.a., and some amplitude modulation can be demonstrated. Also the audible and visible effect of varying the X and Y “offset adjust” can be observed.
Let VA1 be a 1kHz sine wave and VC1 be a l00Hz triangular wave. By varying the bias control the product can be made to rise or fall above the reference level horizon (Fig. 19a). If the sinewave and triangle are produced by VCO1 and VCO2, then the output will also be onesided as both of these signals are one-sided (Fig. 19b). If however, the signal VA1 is alternating, then a double sided output will be produced.
The effects of misalignment of the X and Y “offset adjust” can now be observed. For the Y “offset addust” the result is that the output is non-zero when the control voltage reaches the threshold level. The output may never reach zero or may even pass through zero and become inverted (Fig. 19c). The dynamic range of the v.c.a. is thus severely limited by errors in the setting of the Y “offset adjust” . Misalignment of the X “offset adjust”, results in the unwanted components of VC1, VC2 and the bias appearing at the output, this being particularly disturbing when VA1, is zero.
Voltage controlled filter
The v.c.f. is a bandpass filter with presettable Q factor and a variable (and voltage controllable) centre frequency. The circuit diagram of the v.c.f. is given in Fig. 20. The use of a multiplier makes the circuit appear rather complex; an alterative approach is to use f.e.t. modulators. The result would be a reduction in circuit complexity and cost, paid for at the expense of increased distortion and a loss of linearity of the centre frequency with respect to the control voltage.
The circuit operation is as follows. The sum of the bias and the input control voltage is squared and used to drive the multiplier. The transfer function of the square law generator is given in Fig.21, transistor Tr2 being designed to saturate when its collector reaches a potential of – 5V.
The multiplier is similar to VCA1 and VCA2, but in this case forms part of the main loop with the two integrators. Integrator gains of 3.3 x 10E4 are used and the first integrator is limited by zener diodes D10 and D11. If the output swing of this integrator were allowed its full range of movement the maximum input of the multiplier would be exceeded, causing the loop to become unstable and then “hung-up”. The Q factor can be modified by adjusting R22 and theoretically when the wiper of this pot is at 0V there is no damping term, and the loop becomes unstable. This situation may or may not occur it being dependent on component imperfections and the way in which the circuit is constructed.
It was decided that the filter should not become oscillatory at high Q factor settingd. Thus to eliminate this possible state, R23 has been included, the value of which is chosen to make the loop non-oscillatory throughout its ranges. The v.c.f can be modified externally so that it forms a low distonion oscillator, but this will be decscribed later. Other synthesizers tend to prefer the use of low pass v.c.fs, and this is easily achieved by taking the output from IC4 the low pass output. In fact there is no reason why the bandpass and the low-pass outputs should not be simultaneously available.
Careful adjustments of presets R42, R44, and R35 are required, because the multiplier is part of a loop that can easily become unstable, particularly at low values of gain.
Break link A, Fig. 20, and align the multiplier using the method described for VCA1 and VCA2. Note that there is no “gain preset” to adjust. Replace link A andmonitor the output of IC5. ‘This potential may need to be zeroed by adjusting R35, this offset being particularly large when the bias control is set at minimum. If the multiplier has been carefully aligned to give a maximum dynamic range, then a range in centre frequency of about one octave can be expected. Note that the multiplier’s gain should not quite reach zero or (even worse) pass through it and hence change sign, as both of these states are unstable.
The v.c.f. is now functional and can be used to perform a variety of operations. The more common uses are to filter inputs such as white noise, producing various effects similar to wind, rain, jet engines etc.
The v.c.f can be used as a low distortion oscillator (Fig. 22a). The Q factor and the bias are set at maximum and the control pot R18 (Fig. 20) is increased so that the filter just oscillates. The v.c.f. can now be swept throughout its range, producing a sinewave of virtually constant amplitude and of variable frequency (a 9.5 to 1 range was obtained). There is a 90-degree phase difference between thee low-pass and band-pass output, sine and cosine waveforms ˛eing simultaneously available.
The v.c.f. can also be used as a notch filter (Fig. 22b). To set up the mixer, use a sine wave input and set the v.c.f. to the same frequency, with the Q factor at maximum. By varying the pots VA1 och VA2, the input and the bandpass output can be made to cancel out. The “notch” can then be swept throughout the v.c.f’s range. This technique can be used to examine the harmonic content of any signal that lies within the frequency range of the v.c.f.
The v.c.f is also useable as a spectrum analyser, measuring energy per hertz versus frequency (Fig. 22c). However, this is no more than a demonstrarion piece, due to the narrow range. Also, the sweep time must be rather long if a “high” resolution (i.e. high Q factor) is required, and the constant bandwidth makes interpretation of results difficult.
Audio mixer and summer/inverter
The audio mixer is a three channel virtual-earth mixer, each channel having its own attenuator and being a.c. coupled. A master volume control determines the overall signal level at the output (Fig. 23).
Also, two direct coupled virtual-earth mixers are provided (Fig. 24). These both have three inputs, two having a fixed gain of – 1 and one of – 10, and are used for signal processing, such as inversion, summing or amplification.
The reverberation unit consists basically of three sections; the driver, the springline reverberation unit and the equalised pickup amplifier (Fig. 25). The springline reverberation unit used was the “Hi242” obtainable from Henry’s Radio. This unit is moderately inexpensive, but suffers from a loss of high frequency reverberation, dropping considerably at about 4kHz. However, a useful response can be obtained – enough in fact for this unit to be used in one of the commercially available synthesizers.
By operating switch “S” a choice of the input signal plus reverberation, or just reverberation is available. Thus the reverberation can be separately controlled, by using a v.c.a. and/or a v.c.f., as well as being manually controllable (R3). To reduce any microphonic effects, the HR42 unit should be mounted on a pair of rubber pillars.
This unit has an exponential transfer function of the form, Vout = exp( Vin +constant)
The base-emitter junction of transistor Tr3, Fig. 26, is voltage driven, the collector current being monitored. The relationship between Vbe and Ic is very nearly exponential, modified by the fact that the voltage drive is imperfect and the value of Vce (Tr3) is changing. The suitable working range for the base emitter voltage of Tr3 is from 0.5V to about 0.7V, a width of only 200mV. This requires that Tr3 is biased to a Vbe of about 0.5V and that the control voltage drive is suitably attenuated, the bias being preset by adjusting R9. As two exponential converters are included in the synthesizer, both should be adjusted so that their respones are matched.
The need for an exponential transfer function is twofold. One, the subjective response to volume can be loosely described as “logarithmic”. And two, the subjective response to a change in frequency is governed by the ratio ofthe two frequencies. Thus, frequency generation should be an exponential function of the control signal, if the control, from say a set of keyboards, is to be considered musically useful.
The construction of the synthesizer will be completed with a description of the sample and hold function, noise sources, waveform generator and power supply. All the syntbesizer functions will then be linked via the patch panel, keyboards and joystick control. Details of i.c. pin connections will be given and also examples of the
In Fig. 13, a resistor R10 of value lkOhm should be inserted between the + 15V supply and zener D3.
- D. T. Smith, ˛’\˛iultivibrators with Sevendecade Range in Period”, Wrrefess World,
February l972, pp. 85-86.